Presently, modern power, principally operational amplifiers and other amplifiers incorporating a differential pair configuration have an inherent slew rate limit which limits their performance as the output signal approaches an upper frequency limit. This slew rate induces distortion in the amplifier, often designated as slew induced distortion (SID) or transient intermodulation distortion (TIM) throughout the audio industry. If the signal is an audio signal, this slew rate distortion affects the acoustic quality of the output signal. In particular, modern stereo equipments all comprise an amplification stage. The sound quality of such a stereo equipment has been increasingly improved in the past decade. However, the slew rate distortion cruelly limits further improvements which can be brought to the sound quality.
A complete study of slew rate distortion, in particular in connection with audio signals, can be found in an article by Walter G. Jung, et al., "An Overview of SID and TIM," Audio, June 1979. This article is incorporated herein by reference.
Slew rate elimination is an issue which has been addressed in numerous patents and articles. One of the reasons which seems to account for the slew rate distortion of modern amplifiers, in particular of amplifiers presently used in audio equipments, resides in the topological structure of the amplifiers used therein. The person skilled in the art will recognize that most modern operational amplifiers utilize a differential pair in their input stage. Differential pairs are well known in the art and are one of the most elementary building blocks of the analog electronics industry. A differential pair can also be found at the heart of the common flip-flop, an important building block in digital electronics. It has been relied on as a veritable cornerstone throughout the electronics industry.
The differential pair is typically comprised of two transistors or active devices arranged in a mutually exclusive "see-saw" configuration. Applying a signal which turns on the first device, simultaneously turns off the second device and vice versa. It is quite linear around the 50%/50% center balancing region. However, its linearity drops off quickly, and often severely, on either side of the balance region.
Reference is now made to FIG. 1 which is a circuit diagram of a differential pair. A differential pair typically comprises a pair of transistors Q.sub.A and Q.sub.B. The base of the transistor Q.sub.A is connected to a first terminal 1, to which is applied a tension V.sub.A whereas the base of the transistor Q.sub.B is connected to a second terminal 3 to which is applied a tension V.sub.B. The transistors Q.sub.A and Q.sub.B have their emitters connected together at a node 4 to the terminal 5 to which a voltage -V is applied. A voltage V is applied between the two sources of the transistors Q.sub.A and Q.sub.B. A constant current source 7 drives a current I.sub.A through the transistor Q.sub.A and a current I.sub.B through the transistor Q.sub.B, respectively, by means of a current rail 9 and a current rail 11. Two stray capacitances C.sub.1 and C.sub.2, designated by the numerals 13 and 15, are connected between the rails 9, 11 and ground. The total current flowing through node 4 and the terminal 5 designated as I.sub.bias, typically in the order of 1 mA, is simply the sum of I.sub.A and I.sub.B. When the voltage V.sub.A is applied at the terminal, it turns on the transistor Q.sub.A which pulls more current I.sub.A down on the rail 9 because of the high input impedance and current gain of the transistor Q.sub.A. The increase of current I.sub.A provokes a rise in the voltage across the source-emitter of the transistor Q.sub.A. As the base of the other transistor Q.sub.B is still grounded to the terminal 3, the emitter to base voltage of the transistor Q.sub.B is closed off. This property in a differential pair to shut off one transistor when the other transistor is turned on provides a linear balancing action at the center region. The function I.sub.A with respect to the voltage V.sub.A is basically non-linear with the exception of a narrow linear domain at the center and is schematically represented by the graph illustrated in FIG. 2a. The function is mathematically represented by the following equation: EQU I.sub.A =I.sub.o tan h (V.sub.A /V.sub.o) (1)
where V.sub.o and I.sub.o are both constants.
Typically, an amplifier, for example an operational amplifier comprises two stages: an input transconductance amplifier and an integrating amplifier. The transconductance amplifier generally comprises a differential pair. As long as the voltage V.sub.A remains less than a specific upper limit, the transconductance amplification function represented by Equation (1) remains remarkably linear. Usually, this upper limit to V.sub.A is about 50 mV and up to 100 mV in some amplifiers. Beyond this range, either in the positive or in the negative values, the current I.sub.A reaches a saturation plateau, either transistor is driven full on.
Until recently, this small linear region of the differential pair was assumed to be largely adequate for all types of signal processing. However, particular attention should be paid to the saturation regions of the differential pair, where the other transistor is driven completely off.
Referring to FIG. 2a again, when the current I.sub.bias reaches its maximum, for instance in the positive values, the transistor Q.sub.B is off while the transistor Q.sub.A saturates and latches up. No more input to output variations can be obtained in this non-linear region. At this point, slew rate distortion is generated by the constant current and the stray capacitance 13, 15 in this integrating stage of the amplifier. More specifically, the slew rate distortion is revealed by overshoots and ringing in the output signal as the differential pair tries to regain its center balancing point.
Reference is now made to FIG. 2b which illustrates how a high speed step signal is typically output by an amplifier comprising a differential pair. If a high speed step voltage V.sub.A is applied at the terminal 1, for instance, 10 volts, the transistor Q.sub.A is rapidly saturated and the current I.sub.A reaches a constant value, typically in the order of 100 mA. The point corresponding to the value of 10 Volts is designated by P in FIG. 2a. The integrating stage of the amplifier then integrates this constant value of I.sub.A and provides a voltage ramp which increases linearly with time. This ramp is indicated in FIG. 2b by the numeral 20. The inertia of the amplifier causes the voltage ramp to overshoot the actual value of the output voltage and subsequently ringing happens. As the differential pair amplifier settles down and balances, other overshoots may occur until the final output voltage reaches an equilibrium state. These overshoots are part of the slew rate distortion which the goal of the present invention is to eliminate. The integrating stage amplifier can be designed to reduce the portion of the step signal which is not adequately steep as compared with the input signal (trapezoidal edge). However, an improvement in the slope of the rising ramp may induce higher and more ringing overshoots 23. Alternatively, in order to avoid high overshoots, slow amplifiers can be used but they provide very slow, shallow trapezoidal rise times.
In addition to being non-linear for large input signals, the differential pair's saturation region causes serious stability problems. In both the audio and robotics/servo industries, there is a great deal of signal content containing instantaneous transients, which often exceed the linear region of the differential pair by several orders of magnitude. These transients are considered as large signals and generally move faster than the balancing action of the differential pair. These transients drive the differential pair well into saturation for a short, but definite period of time. If the transient is nearly the same speed as the balancing action of the differential pair, a dangerous phase shift can develop between the input and the feedback network, causing "run-away" oscillations.
Feedback compensation for high-frequency stability is generally necessary for both operational amplifiers and typical power amplifiers. The basic theory and practice of compensation for differential pair is centered around the concept of reducing the high-frequency, closed loop gain below unity before the internal phase shift reaches -180. This is where oscillations naturally start increasing if the gain seen at the inputs is greater than or equal to one. Compensation by its very nature severely restricts the high-frequency performance of typical amplifiers. Anyone skilled in the art of prior art operational amplifiers should be well aware of the need for high frequency feedback compensation.
The use of differential pairs in the audio technology has raised numerous problems which have not been yet resolved. The slew rate, which is caused by a saturated differential pair, severely limits the large signal bandwidth (BW) capabilities of typical amplifiers. Thus, the large signal bandwidth can never equal the small signal bandwidth. For this reason, any amplifier which comprises a differential pair configuration, has a dynamic frequency response that is inversely proportional to input signal levels. Thus, the greater the input, the narrower the bandwidth of the amplifier.
This is illustrated in FIG. 3a which shows how the large signal bandwidth is limited by slew rate for audio-signals. In FIG. 3a, the coordinates are respectively the frequency (abscissa in Hertz) and the power of the acoustic signal in decibels. Typically, a maximum steady state sound pressure level is in the order of 100-120 dB in ordinate. However, short impacts may be substantially louder, of the order of 160 dB. The large signals are adequately covered up to 500 Hz. This portion is designated by the numeral 27. The small signal bandwidth, designated by the numeral 28, does not show a slew rate limited response. However, higher frequencies for large signals are inadequately covered due to the slew rate limitations. The slew rate curve portion is indicated by the numeral 29. This causes the amplifier's sonic characteristics to continually change throughout loud musical passages, resulting in the rather harsh or cold metallic sound that typical solid state amplifiers are known for.
Another serious problem arises as signal levels increase. Low level texture and ambience are literally erased, as the differential pair is driven into saturation, or effectively off, for a certain percentage of time during any given crescendo. This causes a unique "comb-like" eraser of the low level signals and ambience during loud musical passages. This explains why concert hall ambience is considered most difficult to record during a live performance. The current electronics being used have erased this ambience long before it reaches the tape machine. The subconscious ear can hear these minute, discontinuous, missing bits of background instruments, but can only translate this information to the conscious brain as stereo, not live music. The conscious ear is not fast enough to notice these missing sections.
Electron tubes have a saturation threshold which is generally 10 times greater than transistors. In addition, the saturation corners are much softer. These types of amplifiers can track larger transients before the effects of saturation are noticed, and the transition into saturation is more gradual. This accounts for the warmer, more realistic, or natural, sound of tube amplifiers, which many musicians still prefer over conventional transistor amplifiers. However, electron tubes are difficult to handle, expensive, fragile and cannot be miniaturized.
There is therefore a need in audio and servo/robotics for amplifiers which can handle complex wideband signals with no slew rate distortion, overshoot, or ringing, comprising inexpensive elements and simple to manufacture.